1. Field of the Invention
The present invention is directed to a discharge lamp driving circuit for operating a discharge lamp by a high frequency alternating current converted from a low frequency alternating current source such as AC mains.
2. Description of the Prior Art
There has been provided a discharge lamp driving circuit for operating a discharge lamp by a high frequency alternating current. Such driving circuit is required to suppress an input current distortion as well as to maintain a high input power factor. For achieving a high power factor, various circuits have been proposed to include a step-up chopper for conversion of an AC voltage source into a DC voltage source and an inverter for conversion of a DC current from the DC voltage source into a high frequency AC current being fed to operate the discharge lamp.
However, such lamp driving circuit having the two-stage conversions at the chopper and the inverter necessitates a relatively large number of electric components, increasing the bulk and cost of the circuit. In order to reduce the bulk and cost, there have been also proposed discharge lamp driving circuits of various configurations.
Japanese Patent Early Publication (KOKAI) No. 4-193067 proposes a circuit having a circuit configuration of FIG. 21 of the attached drawings which is equivalent to that shown in FIG. 6 of the publication. In this circuit, a series combination of diodes D1, D2 and a smoothing capacitor Ce is connected across a full-wave rectifier diode bridge DB to provide a DC power source from an alternating voltage source AC. A series combination of switching elements Q1 and Q2 is connected across the smoothing capacitor Ce. Another series combination of a DC current blocking capacitor Cc, an inductor Lrs, and a capacitor Crs is connected across the one switching element Q2, while a discharge lamp Ld is connected across the capacitor Crs as a load. The switching elements Q1 and Q2 are cooperative with capacitor Cc to form the inverter of a half-bridge configuration and are controlled by a controller (not shown) to alternately turn on and off at a high frequency sufficiently higher than the frequency of the AC power source. MOSFET is utilized as the switching elements Q1 and Q2. Thus configured inverter operates to convert a DC voltage across the smoothing capacitor Ce into the high frequency electric power which is then fed to the discharge lamp Ld through a resonant circuit of capacitor Crs and inductor Lrs. In order to suppress an input current distortion for keeping a high input power factor, a capacitor Cin in included in a path between an output of the inverter (connection of inductor Lrs to capacitor Crs) and a point between diodes D1 and D2.
Now considering a transient operation in a short time (i.e., corresponding roughly to one switching cycle of switching elements Q1 and Q2) of the circuit of FIG. 21, the circuit can be represented as shown in FIG. 22 in which output voltage Vg of rectifier DB is connected to the anode of diode D1, a DC source voltage Vdc is connected to the cathode of diode D2, and a high frequency source voltage Va is connected through a capacitor Cin to a point between diodes D1 and D2. Since the rectifier DB is assumed to give a constant output voltage Vg within one cycle of the high frequency voltage Va, a constant voltage Vdc is developed across smoothing capacitor Ce. In the following description, voltage Va being applied to the discharge lamp is explained to have a amplitude Vp.
The operation of the above circuit can be explained through four successive stages as shown in FIGS. 23A to 23D. FIG. 23A illustrates an operation at one of four stages corresponding a period 1 of FIG. 24 in which voltage Va decreases from a positive peak Vp. In this stage, diodes D1 and D2 are both made non-conductive so that capacitor Cin does not discharge to maintain a voltage Vc across capacitor Cin at a minimum voltage Vc.min. FIG. 24 illustrates a charge-discharge current Cin flowing into and from capacitor Cin. Minimum voltage Vc.min within one cycle of voltage Va corresponds to a difference between voltages Vd and Vp. In this period 1 where capacitor Cin provides a constant voltage, voltage Vb at the connection between diodes D1 and D2 decreases with a decreasing voltage Va. The period 1 continues until voltage Vb at the connection between diodes D1 and D2 decreases to voltage Vg (=Va+Vc.min).
When voltage Vb becomes equal to Vg (=Va+Vc.min), diode D1 is made conductive, as shown in FIG. 23B, to start a period 2 of FIG. 24 where capacitor Cin receives a charging current Ic. Since the voltage source AC has only low impedance, i.e., sufficiently large current capacity, voltage Vb at the connection between diodes D1 and D2 is maintained at Vg, as shown in FIG. 24. That is, voltage Vc across capacitor Cin will increase as voltage Va decreases. When voltage Va reaches a negative peak voltage--Vp, no charge current Ic flows into capacitor Cin to thereby make diode D1 non-conductive, thereby terminating the period 2. At this occurrence, voltage Vc across capacitor Cin increases to a maximum voltage Vc.max within one cycle of voltage Va.
In the subsequent period 3, voltage Va will increase from the negative peak voltage--Vp, as shown in FIG. 24. In this period, diodes D1 and D2 are made both non-conductive, as shown in FIG. 23C, so that capacitor Cin will not discharge to maintain voltage Vc across capacitor Cin constant at a maximum voltage Vc.max, as shown in FIG. 24. That is, voltage Vb between diodes D1 and D2 will increase with the increasing voltage Va. The period 3 will last until voltage Vb is made equal to voltage Vdc (=Va+Vc.max).
When voltage Vb becomes equal to voltage Vdc (=Va+Vc.max), a period 4 appears to make diode D2 conductive, as shown in FIG. 23D.
In this period 4, a discharge current Ic will flow from capacitor Cin through diode D2, as shown in FIG. 24. Since the smoothing capacitor Ce has a sufficiently low impedance (or great capacity), voltage Vb between diodes D1 and D2 is kept at voltage Vdc. That is, as voltage Va increases as shown in FIG. 24, voltage Vc across capacitor Cin will decrease. When voltage Va reaches the positive peak Vp, no further discharge current Ic flows to make diode D2 non-conductive, terminating the period 4. At this occurrence, voltage Vc across capacitor Cin decreases to the negative peak voltage Vc.min so that the period 1 takes over.
As described in the above, the periods 1 to 4 repeat as a consequence of the switching elements Q1 and Q2 being turned on and off, and an input current is fed from the voltage source AC in the period 2. Thus, the voltage source AC can supply a high frequency current while the switching elements Q1 and Q2 are turned on and off such that the provision of high frequency blocking filter between the source AC and rectifier DB enables to continuously flow the input current from the volage source AC for suppressing the input current distortion. Also as is clear from the above operation, the length of each of periods 1 to 4 will vary depending upon the level of the input voltage Vg. For example, while the input voltage Vg is maintained at its peak value (i.e., Vg=Vdc), periods 1 and 3 do not appear so that the length of each period 2 and 4 becomes maximum corresponding to half cycle of voltage Va. As such, the input current will flow in an amount nearly proportional to an absolute value of voltage Vb for maintaining the input power factor at a high level. It is noted here that the forward bias voltage drop of diodes D1 and D2 is neglected in the above explanation, and that resistor R in FIGS. 23A-23D corresponds to the inverter and resonant circuit.
Operation of the resonant circuit as a load of the inverter in the circuit of FIG. 21 will be now discussed. In periods 1 and 3 where diodes D1 and D2 are both made non-conductive, capacitor Cin is excluded from the load of the inverter so that the circuit can be understood as an equivalent circuit of FIG. 25A. Capacitor Cc is selected to be of sufficiently high capacitance not to influence upon a resonant frequency of the resonant circuit. The resonant frequency in these periods is therefore determined by inductor Lrs and capacitor Crs. In periods 2 and 4, one of diodes D1 and D2 is made conductive so that capacitor Cin becomes an additional factor of determining the resonant frequency so that the circuit can be understood as an equivalent circuit of FIG. 25B. Thus, the resonant frequency in these periods is determined by a parallel combination of capacitors Crs and Cin plus inductor Lrs. In this manner, the resonant circuit changes its configuration (hereinafter referred to as resonant mode) within one cycle of voltage Va. Also, as explained in the above, since the length of the periods 1 to 4 will vary in accordance with an instantaneous value of input voltage Vg, an envelop of the lamp current flowing through the discharge lamp Ld within one voltage cycle of the voltage source AC will vary in accordance with the instantaneous value of input voltage Vg. In this consequence, there appears increased ripple and crest factor in the envelop, resulting in undesired fluctuation of light output with associated flickering.
In order to avoid the above problem, U.S. Pat. No. 5,410,466 having the same basic circuit configuration as mentioned in the above proposes to add a control scheme for controlling the operating frequency of the switching elements Q1 and Q2 and duty ratio thereof in order to suppress the crest factor of the lamp current. However, this scheme is designed to suppress the crest factor of the lamp current only during the normal steady-state lamp lighting operation, and cannot do so during a dimmer operation of dimming the lamp for the following reason.
FIG. 26 illustrates individual characteristic curves of output gain at the different resonant modes in the above periods 1 3 and 2 4 in which (a) is for indicating the characteristic curve obtained in the periods 2 4 at the dimmer operation, (b) for the curve obtained in the periods 1 3 at the dimmer operation, (c) for the a curve obtained in the periods 2 4 at the normal lighting operation, and (d) for the curve obtained in the periods 1 3 at the normal lighting operation. A switching frequency can be selected to be .function.0 where curve (c) crosses with curve (d) so as to turn on and off the switching elements Q1 and Q2 for the normal lighting operation of the lamp. Thus selected switching frequency can therefore reduce the variation in the output current due to the changing resonant modes, thereby enabling to suppress the ripples in the lamp current during the normal lighting operation.
A frequency control could be adapted in the above lamp driving circuit including the resonant circuit to vary the switching frequency of the elements Q1 and Q2 in accordance with the input voltage. A control signal utilized in this frequency control has a varying frequency of which bandwidth (i.e., modulation width) is dependent upon the amplitude of the input voltage. Since the amplitude of the input voltage is nearly constant, the modulation width is also kept nearly constant. Therefore, the frequency control is found effective to reduce the ripples and crest factor of the lamp current during the normal lighting operation.
Discussion is made to the dimmer operation which is effected by varying the switching frequency of switching elements Q1 and Q2. For example, when making the dimmer operation by shifting the switching frequency to .function.1 higher than that for the normal lighting operation, there appears a large difference between the output gain (indicated by .box-solid. in FIG. 26) during periods 2 4 and the output gain (indicated by .quadrature. in FIG. 26) during periods 1 3, resulting in a correspondingly large difference in the output current between at the zero-cross point and peak of the input voltage. Even if the above frequency control is added, the modulation width is held constant irrespective of a varying dimming extent. Therefore, the crest factor of the output current is not expected to be improved, and even the operating life of discharge lamp Ld is considerably shortened when making the dimmer operation.
Alternately, a duty control may be utilized to vary a duty ratio of switching elements Q1 and Q2 instead of the switching frequency for effecting the dimmer operation. This control is made at a fixed switching frequency but is accompanied with varying equivalent impedance of the discharge lamp Ld. Consequently, there also appears a large difference between the output gain (indicated by .DELTA. in FIG. 26) during periods 2 4 and the output gain (indicated by .tangle-solidup. in FIG. 26) during periods 1 3, resulting in a correspondingly large difference in the output current between at the zero-cross point and peak of the input voltage. The duty control can be also combined with the above frequency control. However, since the modulation width is held constant irrespective of a varying dimming extent, the crest factor of the output current is not expected to be improved, and even the life of discharge lamp Ld is shortened when making the dimmer operation.
In short, the dimming of the discharge lamp either by the frequency control or duty control results in the increased ripples and crest factor to thereby shorten the life of the discharge lamp.
In the meanwhile, it is known that the discharge lamp will vary its equivalent impedance with a varying environmental temperature. Also when dimming the lamp, the equivalent impedance will increase with a correspondingly reduced lamp current. The increased impedance acts to enlarge the difference between the output gains of the two resonant modes within one cycle of the switching elements Q1 and Q2, thereby further increasing the low frequency ripple. Therefore, when dimming the lamp at a low environmental temperature, the discharge may become unstable to show undesired flickering, stripe shifting, or even lamp extinction . Consequently, the dimming of the lamp may shorten the lamp life and even causes the flickering, or the like undesired phenomena at the low environmental temperature.
FIG. 27 illustrates another prior art discharge lamp driving circuit in which discharge lamp Ld and capacitor Crs are connected across the series combination of switching element Q1 and diode D2, in contrast to the circuit of FIG. 21 in which discharge lamp Ld and capacitor Crs is connected across switching element Q2. Further, capacitor Cim is connected in parallel with diode D2 instead of capacitor Cin in the circuit of FIG. 21 for suppressing input current distortion and maintaining high input power factor. The circuit configuration of FIG. 27 can be expressed as an equivalent circuit of FIG. 28 in which an inverter is recognized to form a high frequency power source providing a current of constant amplitude.
The operation of the circuit of FIG. 28 can be explained in terms of four successive stages within one cycle of output current Ia from the high frequency power source, as is made for the circuit of FIG. 22. When the source voltage Vg is at its peak (i.e., Vg=Vdc), diode D1 is kept conductive over a maximum period within one cycle of the output current Ia, corresponding to one half cycle of the switching elements Q1 and Q2.
In the above circuit configuration, a resonant circuit is established by a series combination of inductor Lrs, capacitor Crs, and capacitor Cim while diodes D1 and D2 are both non-conductive. When diode D2 becomes conductive, capacitor Cim is shunted so that a resonant circuit is established by a series combination of inductor Lrs and capacitor Crs. Thus, this circuit has also two resonant modes within one switching cycle of switching elements Q1 and Q2, as is seen in the circuit of FIG. 21, and therefore gives rise to the same problem that the envelop of the lamp current will vary with the input voltage Vg to have increased ripples with attendant increase in the crest factor, thereby shortening the lamp life.
It has been also proposed in U.S. Pat. No. 5,404,082 and No. 5,410,221 to control the switching frequency of the switching elements Q1 and Q2 in the circuit of the like configuration for reducing the crest factor of the lamp current. The control is made to detect the input voltage, output voltage and lamp current so as to vary the switching frequency in accordance with detected parameters for reducing the crest factor of the lamp current. However, this prior art circuit is found to suffer also from increased crest factor at the time of dimming the lamp.
That is, the circuit of U.S. Pat. No. 5,404,082 operates to control the switching frequency based upon the detected input voltage, and suffers from varying ripple and crest factor with varying extent of dimming the lamp, as explained hereinbefore with reference to FIG. 26.
The circuit of U.S. Pat. No. 5,410,221 is designed to vary the switching frequency based upon the detected output voltage to the discharge lamp Ld for reducing the crest factor. In this circuit, a control is made to give a constant ratio between amplitude of variation in the lamp current and modulation width of the frequency of the switching elements Q1 and Q2. When dimming the lamp with the use of thus configured circuit, the ripple will become greater while the lamp current is made small. Therefore, the control signal is unable to give a modulation width wide enough to remove the ripple, eventually failing to reduce the ripple to a satisfactory extent at the time of dimming the lamp and suffering from increased power factor, thereby leading to unstable light output and shortening of the lamp life.
A further prior art circuit has been proposed by the inventors of the present application in the paper entitled "An Improved Charge Pump Electronic Ballast with Low THD and Low Crest factor" published by IEEE APEC '96 Conference Proceedings, pp. 622-627, 1996. As shown in FIG. 29, the circuit comprises a full-wave rectifier DB composed of a diode bridge for full-wave rectification of an alternating current voltage source AC such as AC mains, a smoothing capacitor Ce connected through a diode D2 across the outputs of the rectifier DB, and a pair of switching elements Q1 and Q2 connected in series across the smoothing capacitor Ce. A series combination of an inductor Lrs and capacitor Crs is connected across switching element Q2 on negative terminal side of smoothing capacitor Ce. A series combination of an inductor L2 and a capacitor C2 is connected across capacitor Crs through a DC blocking capacitor Cc. A discharge lamp Ld is connected across capacitor C2. Also, a diode DC1 connected between one end of inductor Lrs adjacent to capacitor Crs and the anode of diode D2. Further, a diode DC1 is connected between one end of inductor Lrs adjacent capacitor Crs and the cathode of diode D2 with the cathode of diode DC1 connected to cathode of diode D2. Connected across capacitor Crs is a diode DC2 having its anode connected to negative terminal side of the rectifier DB. With this configuration, the circuit has two resonant circuits, one composed of inductor Lrs and Crs and the other of inductor L2 and capacitor C2.
The circuit of FIG. 29 prevents smoothing capacitor Ce from having increased voltage Vdc at a light load operating condition such as pre-heating or starting-up of the lamp, thereby avoiding undue voltage stress which would otherwise applied to circuit components. Diodes DC1 and DC2 are provided to suppress the crest factor. Diodes DC1 and DC2 act to clamp the peak-to-peak voltage across capacitor Crs to voltage Vdc across smoothing capacitor Crs to keep voltage across capacitor Crs clamped at voltage Vdc across smoothing capacitor Ce. Thus, the input voltage to the resonant circuit of inductor L2 and capacitor C2 is made to have a constant amplitude, thereby reducing ripple and therefore crest factor of the lamp current being fed to the discharge lamp Ld. Also because of that the peak-to-peak voltage across capacitor Crs is restricted to voltage Vdc of smoothing capacitor Ce, the envelop of the voltage being applied to capacitor Cin takes a sinusoidal form in conformity with the input voltage, thereby reducing input current distortion. This is confirmed from waveform comparison between FIGS. 30A, 30B in which diode DC1 and DC2 are eliminated and FIGS. 31A, 31B in which diodes are included. FIG. 30A and FIG. 31A show waveforms of voltage across capacitor Crs, while FIG. 30B and FIG. 31B show waveforms of voltage across capacitor Cin. In these figures, Vdc and Vg indicate voltage across smoothing capacitor Ce and output voltage of rectifier DB, respectively.
As explained in the above, the circuit of FIG. 29 is contemplated to suppress the crest factor of the lamp current without relying upon the frequency control of the switching elements Q1 and Q2. However, when dimming the lamp by a duty control of varying duty ratio of switching elements, there appears the following problem. The duty control is made to give the normal lighting operation at the duty ratio of 50%, i.e., at 1:1 ratio between ON-time duration of switching element Q1 and that of switching element Q2, and to give the dimming operation at a varying ON-time ratio between switching elements Q1 and Q2. For example, when the ON-time ratio is 7:3 between switching elements Q1 and Q2, voltage variation across capacitor Crs is reduced in its amplitude to thereby reduce the current flowing through capacitor Cin from voltage source AC, thereby lowering both the input from voltage source AC and output voltage to the discharge lamp Ld and maintaining a constant voltage Vdc across smoothing capacitor Ce.
However, the dimming of the lamp involves the reduction of voltage across capacitor Crs, while voltage Vdc across smoothing capacitor Ce is kept constant. This means that voltage across capacitor Crs is not clamped, thereby increasing the crest value of voltage across capacitor Crs being fed as input voltage to the resonant circuit of inductor L2 and capacitor C2 and therefore increasing the crest factor of the lamp current being fed to the discharge lamp Ld.
In order to suppress the crest factor of the lamp current, a modification may be conceived as shown in FIG. 32 in which a current sensor SI in the form of a current transformer is provided to detect the lamp current and a control is made to vary the operating frequency of the switching elements Q1 and Q2 based upon the detected lamp current. For this purpose, a feedback circuit FB is provided to include an error amplifier Amp and an delay circuit of resistor R1, diode Da, and capacitor Cd. The lamp current detected at the current sensor SI is converted into a corresponding voltage by means of resistor Rd, and is then processed in the delay circuit to give the ripple in an envelop of the lamp current which is compared with a reference voltage Vref to give a resulting error therebetween to a control circuit CN. The control circuit CN responds to vary the frequency of a control signal from the control circuit CN in a direction of eliminating the error. With this configuration, the lamp current can have a reduced crest factor at the rated lamp lighting.
However, when intended to apply a dimmer signal Dim to the control circuit CN for dimming the lamp, as shown in FIG. 33, there appears a problem that the crest factor of the lamp current will increase. This is because that the dimming of the lamp reduces the lamp current to correspondingly reduce the current fed to the feedback circuit FB. As discussed hereinbefore, voltage across capacitor Crs is not effectively clamped by diodes DC1 and DC2 while dimming the lamp so that only small output of the feedback circuit FB is available while the lamp current sees a large variation. Consequently, such lamp current of reduced level but of large variation is insufficient to modulate the control signal in a predetermined range given to the control circuit CN, thus failing to compensate for the large variation in the lamp current and therefore failing to reduce the crest factor successfully.
Notwithstanding that the above prior art discharge lamp driving circuit of a charge-pump type in which capacitors Cin and Cim are interposed in a charging path between the output of the inverter and smoothing capacitor can reduce the ripple and the crest factor of the lamp factor in the normal lighting operation, the circuit had to suffer from increased ripple and crest factor when dimming the lamp. In addition, the lamp current will suffer from a large variation at a low environmental temperature to bring about the undesired flickering.